User:Daeyun Kim/sandbox/2018s2-297 Wireless Rotation Detector


 * 1) Supervisors
 * 2) Masters students
 * 3) Project objects
 * 4) Scope
 * 5) Aims and motivations
 * 6) Background & Significance
 * 7) Proposed Approaches
 * 8) Numbered list item
 * 9) Key Requirements
 * 10) Technical Challenge
 * 11) Proposed Approaches
 * 12) Schedule and Milestone
 * 13) Metamaterial Absorber Design
 * 14) Antenna Design
 * 15) Software Design
 * 16) Entire System Test
 * 17) Conclusion
 * 18) Reference

Projects:2018s2-297 Wireless Rotation Detector

Supervisors Prof Christophe Fumeaux Dr Jammy. shengjian chen

Honours students Daeyun Kim(a1730642) Yafei Zhao

Project objects Designing and constructing a portable device which can measure the speed of a rotating device by using Microwave Improving the current design performance by installation of superstrate over the transmitter and receive antennas Making reflectors which can detect 180 ambiguity of polarization

Scope a portable wireless device which has transmitter and receiver reflectors (absorbers) software to measure the speed of rotating reflectors

Specific Aims

The aim of the project is to design and construct a portable wireless device which can be used for rotation speed measurement. The device is supposed to consist of antennas, absorbers, Arduino UNO Board with software, etc. The outcomes of the project will be as follows: A metamaterial absorber with resonant frequency of 2.4GHz A planar dual-antennas with capability of transmitting and receiving signals A software system for rotation speed calculation

The device should be able to measure the rotational speed of various objects such as sport equipment, car tyres, turbines, fans, etc. The system shall be operational under a set of circumstances and able to achieve reliable results.

Background & Significance

1. Polarization of Antennas

2. Polarization Loss Factor

3. Metamaterial Absorber

Key Requirements

Technical Challenge Antenna Design

The technical challenge for antenna design is to figure out an appropriate method to reduce crosstalk between dual antennas with the frequency and substrate thickness specified. Some possible methods have already been mentioned in Section 3.2.2. However, all the methods will have side effects on the performance of the antenna including input impedance matching, efficiency, resonant frequency, etc. Hence, the method should be operational for coupling reduction with all side effects minimized. Furthermore, the method should not dramatically change the overall dimensions of the antenna, and specifically the antenna system is expected to be planar. Metamaterial Absorber

Resonant frequency and absorption ability are two key characteristics of the metamaterial absorber, which are influenced by the metallic structure and thickness of substrate. Hence, 2 variables need to be considered in absorber design in order to achieve the requirements. Moreover, the material of substrate is supposed to be FR4, a low-cost material with permittivity varying from 4.2 to 4.8. The variation of the permittivity has impact on the resonant frequency, which may lead to failure in absorber design. Hence, a relatively wide bandwidth has to be achieved in order to overcome the uncertainty.

Proposed Approaches Method exploited - ‘Divide-And-Conquer’. The entire project can be decomposed into several aspects. Hardware (design required) 1.Metamaterial Absorber 2.Dual-Antennas Software 1.Calculation Algorithm for rotating speed 2.GUI Other Components (from market or school) 1.VCO 2.Arduino UNO Board 3.Power Detector

Hardware Design

Hardware design will begin with understanding the key requirements and specifications of the hardware. Most of the hardware design will be implemented in the commercial software package HFSS which is a professional software for simulating 3D full wave electromagnetic fields. The group member will fabricate the designed hardware after hardware has been designed in HFSS. The next task for hardware is to measure the prototype and compare the practical result with simulation results. If the prototype fails to meet the requirements, the group will refer back to design stage to redesign prototype until it meets requirements

Software Design

The software design will start after antenna and absorber design have been finished. The software design will be in C language based on the Fourier Transform. To test the validity of the software, the group members will combine software, hardware and other components together. If the software cannot accurately measure rotating speed of objects which is driven by a motor, the project group will return to previous stage to redesign the software.

Schedule and Milestone Milestone in Semester 1

Table 1.Milestone in SemesterMilestone in Semester 2

Table 2.Milestone in Semester 2 Metamaterial Absorber Design

1.Method

1.1 Metamaterial Absorber Design

According to the introduction given in the Section 1.3, the existing absorber can only absorb 36% of incident power at 2.4 GHz when it has the same polarization with the antenna. The main cause for this is that the bandwidth of the existing metamaterial absorber is only 0.03 GHz, which is too narrow. As a consequence, the performance of the absorber will significantly drop if the resonant frequency slightly shifts. Accordingly, the main purpose for the new absorber design is to achieve distinct absorption performance and relatively wide bandwidth. This section will demonstrate some different methods which have been tried to increase the bandwidth of the absorber.

1.1.1 Structure Tapering

In section 3.13, we presented the characteristics of ELC resonator and its equivalent circuit. The ELC resonator can be modelled as a parallel LC resonator and the resonant frequency can be adjusted by changing resonator’s dimensions. In order to enhance the bandwidth, we need to consider the impact of the Q factor. For a parallel RLC circuit, the Q-factor is defined as

Q=R√(C/L)

And the bandwidth can be expressed as                                            BW=f_r/Q

In order to reduce the Q factor and remain the same resonant frequency, we tapered the loop width to increase the inductance and increased the gap to decrease the capacitance of the ELC resonator. Figure 4 compares the original ELC resonator and the tapered one. The original structure we proposed has an ELC resonator and a ground plane, separated by the FR4 substrate with height of 3.2mm. The ELC resonator has the dimensions of a=15mm, g=0.4mm, d=12.6mm, b=0.8mm, l=3.6mm and copper thickness of t=0.035mm. The tapered resonator has the gap size varied from 0.4mm to 2.0mm and loop width decreased from 0.8mm to 0.1mm.

Figure 3.ELC_resonator_and_tapered_one From Figure 5 illustrates the reflection coefficient S11 for the metamaterial absorber. The transmission coefficient is 0 because of the metal ground plane. It is interesting to note that the metamaterial absorber resonates at frequency of 2.4GHz with minimum reflection coefficient -18.75dB with the bandwidth is 0.0389GHz.

Figure 4.Reflection coefficient for the metamaterial absorber

Figure 5.Reflection coefficients for tapered MMA

Figure 5 shows the changes in the bandwidth as the gap size increases from 0.4 to 2mm. We can conclude that the bandwidth increases as gap size g increases until 0.8mm, then the bandwidth keeps decreasing as gap size increases, which is completely different from the analysis about Q factor before. In addition, we can observe that the return loss at resonance also increases as the gap size increases. The reason for the results mentioned above can be explained by using interference theory, which leads to the multiple reflection and transmission inside of the absorber structure. Figure 2 shows that the metamaterial absorber has 2 layers: resonator and ground plane. The overall reflection is the superposition of the multiple transmissions and reflections [12], which is defined as:

Formula 1.PNG

where φ1=∅12+∅21+∅23+2β, φ2=∅12+∅23+∅21+2β, β=-√(ε_FR4 ) k_0 d/cos⁡(α_s), k0 is the free space wave number. The formula demonstrates that the overall reflection is associated with the reflection and transmission coefficient of ELC resonator, the reflection coefficient of ground plane, the height of the substrate and the angle of incidence. Hence, the overall bandwidth of the reflection coefficient may be too small if we substitute all relevant values into the formula. Also, due to the complexity of the overall reflection, the bandwidth enhancement by structure tapering may not work for the metamaterial absorber since we only increased the bandwidth of the ELC resonator.

1.1.2 Substrate Height

We then investigated the impacts of the substrate height on the absorptivity and bandwidth. Based on the structure proposed in Section 4.1.1, we varied the substrate height from 2mm to 3.6mm.

Table 3. Bandwidth for different substrate height Table 3 shows the bandwidth as the substrate height changes. It is interesting to note that the bandwidth starts with 0.0164 at 2.4GHz and increases until height increases to 2.8mm. Then the bandwidth start to decrease as the substrate height keeps increases. The optimized result we can obtain was 0.0398GHz, which is still relatively narrow. The reason for this can still be explained by the formula in section 4.11. Due to the complexity of the overall reflection, the bandwidth cannot be significantly improved by only changing the substitute height. In conclusion, the bandwidth still does not meet the specification although it can be slightly increased by varying the substrate height.

1.1.3 Via

After some unsuccessful trials, we finally presented the design of the metamaterial absorber operating at 2.4 GHz by using a via. A unit cell of proposed absorber is shown in Figure 7. It is a 3 layer structure which has a metallic loop at the top, a ground plane and a substrate in between. A 150Ω resistive load is used as a via to connect the top and bottom layer. The absorber is designed to operate at 2.4 GHz with the optimized dimensions given by a1=20mm, a=16mm, b1=19mm, b=13.4mm, w1=4, w2=2.6mm.

Figure 6a.A schematic showing the proposed unit cell of metamaterial absorber

Figure 6b.Prototype of absorber design The unit cell was hosted on the top of a 3.2mm thick FR-4 dielectric substrate with a loss tangent of 0.025 and a dielectric constant of εr=4.4. The simulation for ELC resonator was performed with a full-wave electromagnetic simulator, Ansoft HFSS. The unit cell was placed in a waveguide with Master and Slave boundary condition. The boundary condition was selected for the unit cell in order to realize TEM mode excitation with polarization shown in Figure 7.

Figure 7.Reflection coefficient for the metamaterial absorber

Figure 7 illustrates the performance of the absorber for both co-polarized direction and cross-polarized direction. From the graph, it is obvious that the absorber resonates at 2.4 GHz with reflection coefficient S11 of -26.3 dB for the co-polarization direction. In the meanwhile, the reflection coefficient for cross-polarization is 0 dB at 2.4 GHz. Using scattering parameters, the absorptivity is calculated by A=1-|S11|^2-〖|S21|〗^2, where S21 is 0 due to the ground plane. Therefore, the absorption rate is 99.77% based on the simulation results. In addition, it is interesting to note that the bandwidth of the metamaterial absorber has been significantly increased to 0.2 GHz, which is 7 times more than that of the original design.

Figure 8.Surface current distribution for co-polarization The reason why the design can successfully absorb power can be analysed by the surface current distribution, as shown in Figure 8. From the graph, we can observe that the surface current is mainly distributed at the middle of the left and right side of metallic loop. Meanwhile, the surface current at the top and bottom is minimal. However, the voltage at the surface will be distributed conversely. The voltage at top and bottom of the metallic loop is maximum while that at middle of left and right side is minimum. Therefore, connecting the top layer and ground plane with a resistive load enables the current to flow through the resistor due to the voltage difference, resulting in the power dissipation.

2.Experiment & Result

2.1 Absorber Test

Based on the simulated unit cell structure, a 10×10 metamaterial absorber array has been fabricated, as shown in Appendix B. The unit cells were hosted on the FR4 substrate with height of 3.2mm. The copper thickness for top and bottom layers is 0.035mm. The experiment was conducted in the Anechoic Chamber in order to mitigate the interference from the environment. Figure 11 demonstrates how the experiment was conducted in the Anechoic Chamber. To measure the absorption capability of the absorber, a pair of horn antennas, severing as transmitter and receiver of the power, was placed separately by the distance of 40 cm and connected to the Agilent N5230A analyser with low loss cables.

Figure 9.A schematic for absorber experiment

The absorber was placed in middle front of the 2 antennas. In order to ensure the accuracy of the measurement, the distance between the absorber and the antennas was 140 cm to avoid the near-field effects. However, the transmitting and receiving antennas would have incident angle of 8.13º with respect to the surface of the absorber plane due to the experiment limitations.

Firstly, we measured the power reflected by a copper sheet which has the same size as the absorber. Then, we tested reflection response by substituting the absorber for the copper sheet. The difference between two power responses can be considered as the absorption rate of the metamaterial absorber. The measured results are shown below:

Figure 10.Simulated and measured reflection coefficient for co-polarization

Figure 11.Simulated and measured reflection coefficient for cross-polarization

Table 4.Summary for simulation and measurment for co-polarization Table 4 summarises the simulated and experimental results for the absorber at co-polarized direction. It is obvious that bandwidth, reflection coefficient and absorptivity for both are similar to each other. However, it is interesting to note that the resonant frequency for the measurement is 2.52 GHz, which is different from the simulated one.

Figure 11 shows the reflection coefficient for the cross-polarized direction. We can observe that the reflection coefficient for both simulation and measurement is 0 at 2.4 GHz, which means the absorber can reflect 100% incident power when cross-polarized. In addition, both results show that the resonant frequency at around 3.25 GHz. The difference between simulated and measured minimum reflect coefficients may result from the unwanted edge scattering caused by the finite array of unit cell

Discussion

From the results related to the absorber, we can conclude that the simulated results are very close to the measured ones except for the resonant frequency. The simulated resonant frequency is 2.4 GHz while the measured one is actually 2.52 GHz. This section will include a detailed discussion on the discrepancy between 2 resonant frequencies. The final design uses a 150 ohm resistor as a via to connect the top and bottom layer. In the fabrication process, the resistor we used is called MMA 0204, which is one of the professional thin film MELF resistors.

Figure 12.RF behaviour of MMA 0204 The RF behaviour of the MMA 0204 is illustrated in Figure 12. It is clear that MMA 0204 is purely resistive until 0.6 GHz. After that, the ratio of the total impedance to the resistance significantly increases beyond 1, which means the resistor will have capacitance or inductance at high frequency. From the graph, we can find that |Z|/R is roughly 1.7 at frequency of 2.4 GHz. Hence, the total impedance Z of the resistor at 2.4 GHz is given by: Z=R(1+1.375j), where R is 150Ω

Accordingly, we need to consider the RF behaviour of MMA 0204 at 2.4 GHz in new simulation to obtain more accurate results.

Figure 13.Simulated S11 with RF behaviour of the resistor

Table 5.Summary for simulation and measurement Figure 13 shows the simulated S11 with RF behaviour considered and all results are summarised in Table 5. We can clearly see that resonant frequencies for the simulation and measurement are very close to each other, which means the resistor could be the potential reason which causes the resonant frequency shift.

Antenna Design

1.Method

This section is related to antenna design method and will introduce two failed methods and a successful method that is able to reduce the coupling between two antennas less than -40dB. The absorber design method and software algorithm will be introduced in latter section.

1.1 Antenna structure design method

According to the project requirement, transmission and receiving antennas should be on same side. So the dual patch antennas are decided for this project. The dual patch antennas should be built on same substrate and one is used to receiving and another one is used to transmission. So the antennas structure should be designed at first. The dimensions of the microstrip path antenna are determined by following parameters: Dielectric constant of substrate (εr) Thickness of substrate (h) Resonance frequency of antenna (f)

Based on the three parameters above, the dimension of microstrip patch antenna can be determined. So there are four key parameters are calculated and used for simulation in HFSS. The length (L) of the patch The width (W) of the patch The width (W0) of the feed line The length of rectangle (y0) which should be substrate on patch

1.2 Reducing coupling method

Using the ‘U’ shape structure between two patch antennas. From figure 6, this method is also cutting slots on ground plane between two same directional patch antennas and the ‘U’ shape is composed of three slots. The principle of the ‘U’ shape structure is based on an isolation of reverse phase coupling for reducing coupling.

Figure 14.Dual patch antennas with 'U' structure 2. Antenna results

2.1 Design calculation result:

As referred above, in order to increase the bandwidth of antenna, the RT/duroid 5880 has been chosen as substrate. The substrate parameters and the desired resonance frequency of the antenna are as follow: Dielectric constant of substrate εr is 2.2 Thickness of substrate h is 1.6mm Resonance frequency f is 2.4GHz

According to the calculation methods from the Balanis, the dimension of microstrip line patch antenna can be determined based on above the parameters. The length (L) of the patch The width (W) of the patch The width (W0) of the feed line The length of rectangle (y0) which should be substrate on patch

Table 6.Calculated parameters of patch antenna

Table 7.Optimal result for parameters

2.2 Simulated and measured result

This section includes single antenna, dual patch antennas without ‘U’ structure, and dual patch antennas with ‘U’ structure. Each situation contains simulated result, measured result and comparison as well as relevant discussion. Single patch antenna:

The Figure below illustrates the reflection coefficient at resonate frequency 2.4GHz. The S11 is around -23.78dB at resonance frequency 2.4GHz, which means nearly 99.3% energy has been transmitted through the patch antenna. The reflection coefficient is lower and the input impedance match the transmission line. So the input impedance is close to 50 ohms.

Figure 15.Reflection coefficient for single patch antenna Dual patch antennas (Without ‘U’ structure):

Due to large power drop by using single antenna receiving and transmission, the dual patch antenna is introduced. In the dual patch antennas system, one antenna is used to receiving and another one is used to transmission simultaneously. The following figures illustrate simulated results of reflection coefficient S11 and coupling effect S21 of dual patch antenna without ‘U’ structure.

Simulated result:

Figure 16.Reflection coefficient & coupling effect for dual patch antennas Measured results:

The following figures illustrate measured results of reflection coefficient S11 and coupling effect S21 for dual patch antennas.

Figure 17.Measured reflection coefficient for dual patch antennas

Figure 18.Measured coupling effect for dual patch antennas Comparison:

From figures above, comparing simulated result of reflection coefficient S11 that is around -19dB at resonance frequency 2.4GHz with measured result of reflection coefficient S11 that is -18dB at 2.4GHz, it is obviously that the simulated result is very close to measure results and the resonate frequency is in ISM frequency bandwidth. The S11 result is acceptable. The most important characteristic for the project which should be improved is crosstalk. From figures above, comparing the simulated result of crosstalk coefficient S21 that is around -19dB at resonate frequency 2.4GHZ with measured one which is around -23dB at resonate frequency. The coupling between two patch antennas is too large for the project to accept since it will have large effect on receiving.

Dual patch antennas with ‘U’ structure:

Using ‘U’ structure between two antennas, the following figures illustrate simulated result of reflection coefficient and coupling effect as well as measured. The figure 20 and figure 21 illustrate prototype of dual patch antennas with ‘U’ structure

Figure 19.Simulated results with measured results

Figure 20.Top view of dual antennas

Figure 21.Bottom view of dual antennas Comparison:

From figure 19, after using ‘U’ structure between two patch antennas, the reflection coefficient of simulated is around -19dB at resonate frequency 2.4GHz and measured reflection coefficient is around -24dB at 2.4GHz. Those two results are similar with each other and the performance is acceptable. For the crosstalk efficient, the crosstalk efficient of simulated at resonate frequency is around -43.3dB and the measured crosstalk efficient is -45dB at 2.4GHz. Those two result are close to each other and it is obviously that the coupling between two patch antennas has been reduced dramatically compared with the without ‘U’ structure case. Also from the figure 15, the simulated curve of reflection coefficient overlap the measured one and crosstalk curve of simulated and measured are also similar with each one. The performance of dual patch antennas has met the requirement. The details of principle of ‘U’ will be discussed in next section.

Figure 22.Radiation Pattern for dual antennas in E-plane

Figure 23.Radiation Pattern for dual antennas in H-plane

Figure 24.Radiation Pattern for dual antennas

From figure 22 to figure 24, they demonstrate measured radiation pattern for two patch antennas with ‘U’ structure. All the radiation patterns were measured in the echoic chamber. The two patch antennas system was measured from -180 degree to 180 degree and the measured frequency at 2.4GHz with 1601 sweep points. From the figure 23, it is obviously that the maximum gain in E-plane and H-plane are at 0 degree. So the performance is acceptable and this design has satisfied with the requirement of the project.

Results summary:

Single patch antenna:

Table 8.Simulated and measured result for single patch antenna Dual patch antennas without ‘U’ structure:

Table 9.Simulated and measured results for dual patch antennas without 'U' Dual patch antennas with ‘U’ structure:

Table 10.Simulated and measured results for dual patch antennas with 'U'

Discussion:

This section will focus on some physical explanations about the ‘U’ structure and explain the reason why the ‘U’ structure can reduce the coupling between two patch antennas.

Figure 25.File:Simulated and Measured S21 for dual antennas in different conditions

Figure 26.Schematic for ‘U’ structure

The most important target for this project is reducing the crosstalk between two patch antennas and keeping the antenna system in same plane. Comparing the dual antennas system with ‘U’ structure with antenna system without cut, the figure 25 clearly demonstrates that the coupling between two patch antennas has been reduced nearly 20dB at resonate frequency 2.4GHz no matter what simulation or measurement. The final measured result of coupling is around -45dB which is satisfied with requirement of project. Actually there are a little bit errors between simulated S21 without ‘U’ structure and measured S21 without ‘U’ structure. The reason is the actual patch antennas system without ‘U’ structure was not fabricated. In order to realize this case, the project group stuck some copper slots to cover ‘U’ structure. So glue could a factor which can affect the measured results. However the errors are in a reasonable range so it is still acceptable. As the figure 26 shows, the letter ‘a’ and ‘b’ represent height and width of a slot. Currently the best size of slot can be obtained through HFSS when a is equal to 46mm and b is equal to 2mm. The principle of ‘U’ structure is based on an isolation technology of reverse phase coupling. Assuming the signal comes into left patch antenna. From the figure 25, the black arrow indicates the surface current direction in left antenna. Due to the interference between two antennas, this surface current in left antenna would effect on right one and cause a same direction surface current in right patch antenna. However if there is a ‘U’ structure between patch antennas, from the figure 25 the red arrows indicate the surface current flow direction in slots. It is clearly to see the current flowing in right arm is opposite direction compared with original surface current. This opposite direction surface current would cancel out the original coupling in right antenna. So that is the reason why the ‘U’ structure can be used to reduce coupling for strongly coupled antennas.

Software Design

The software design consists of 2 main parts. One is the calculation algorithm, which is responsible for calculating the rotation speed. The other one is the user interface, which is used to display the measured data to users. The main principle used for the calculation algorithm is Fast Fourier Transform (FFT). According to the discussion in Section 1.2, the reflected power shall periodically vary from 0 to maximum due to the polarization loss. Therefore, FFT transforms the received power in time-domain to frequency-domain and the fundamental frequency of the frequency-domain signal provides the information of the rotation speed. The calculation algorithm is reused from the previous design.

Figure 27.16×2 LCD keypad shield for Arduino By contrast, the user interface is adapted this year to display different information. A 16×2 bit LCD keypad is connected to the Arduino UNO board. From Figure 27, it is interesting to note that there are 6 buttons available in the LCD keypad shield. However, only Left, Top, Bottom and Right are used for the message display.

The functionality of the user interface is listed as follows: The Greeting Message will be displayed on the screen at the beginning. The message begins with “Hi, Welcome” and “Group 28 Wireless Rotation Detector”, which shifts from right to left on the screen, then the students’ and supervisors’ information shows up. The display message can be skipped if “Left” button is pressed. The user interface will then display the rotation speed of objects if the greeting message display is skipped or ends. Pressing “Down” button at speed display enables user to read the information regarding power level. The LCD display will show the maximum and minimum power at the screen in dBm. Pressing “Up” button at Power display will make software display the rotation speed again. The “Right” is used to restart the software system.

Entire System Test

The system test was performed after the absorber and antenna test have been completed. Figure 23 shows the experiment for the system test. We attached the absorber on the centre of a wheel driven by the DC power supply. The dual antennas were hosted at the front part of the housing while the remaining components were mounted on the base.

Figure 23.Pictures showing the system test

The purpose of the system test is to test the performance of the entire system from 2 aspects including accuracy and operating distance. In order to measure the accuracy of the system, a tachometer has been used as a reference. We randomly picked 10 cases which had different rotation speeds, then we measured the speed by using the tachometer and the system we constructed. Finally, we compared the difference between 2 measurements to determine if the system was accurate enough. For the operational distance test, we made the wheel spin at a fixed speed, then we gradually moved the antennas away from the absorber until “No Connection” was displayed on the screen. Meanwhile, the distance between the antenna and absorber could be considered as the maximum operational distance.

Table 11.Measured results from tachometer and constructed system

Table 12.Operating distance for the constructed system Table 11 shows the results measured by the tachometer and the constructed system. It is clear that the results are very close to each other with the maximum error only 3.3%, which means the system has satisfied the requirement of accuracy.

Table 12 provides the information about the operational distance. From the table, it can be concluded that the maximum distance for the system is roughly 70 cm. Given that the operating distance for the original system was 57 cm [2], the system has already been improved from the respective of the operating distance.

Conclusion

The thesis provides the information about the introduction, system design, experiment and result analysis of the wireless rotation detector project. The main objective of this year project is to enhance the performance of the absorber and modify the structure of antenna. For the absorber design, the metamaterial absorber has the resonant frequency at 2.52 GHz with 99.68% absorptivity. The bandwidth of the absorber is 0.17 GHz, which has been significantly enhanced compared to the simulated result (0.03 GHz) from last year. The potential cause for the resonant frequency shift is also discussed in section 6.1, so current structure of the absorber can be modified with the consideration of resistors’ RF behaviour in order to achieve 2.4 GHz resonant frequency. Overall, the absorber design has successfully met the specifications we proposed at the beginning of the project.

On the part of antenna design, the performance of dual patch antennas is acceptable for our project. The reflection coefficient of dual patch antennas is around a low value at resonate frequency and ensure more than 90% power can be transmitted. The challenge for antenna design is keeping the antenna system in a plane and reducing the coupling less than -40dB. So the isolation technology of a reverse phase coupling has been introduced. So this technology has reduced the coupling between two patch antennas less than -40dB, which can ensure that the transmitting signal would not affect the reading.

The requirements about the accuracy and the operating distance of the entire system have also been reached. The entire system can measure the rotation speed within 4% error compared to the commercial product, Tachometer. In addition, the operating distance is also increased from 57 cm to 70 cm under test environment. Overall, the objectives of the project have been achieved and the entire system could be useful for future industrial applications.